Analog to digital converter with current steering stage

ABSTRACT

An analog-to-digital converter (ADC) includes a first ADC stage with a first sub-ADC stage configured to output a first digital value corresponding to an analog input voltage. A current steering DAC stage is configured to convert the analog input voltage and the first digital value to respective first and second current signals, determine a residue current signal representing a difference between the first current signal and the second current signal in the current domain, and convert the residue current signal to an analog residual voltage signal. A second ADC stage is coupled to the first ADC stage to receive the analog residual voltage signal, and convert the analog residue voltage signal to a second digital value. An alignment and digital error correction stage is configured to combine the first and the second digital values into a digital output voltage.

BACKGROUND

Analog-to-digital converters (ADC or A/D) are used in a variety ofapplications in order to convert a sampled analog signal into a digitalsignal. There are a variety of ADC architectures, such as pipelined,flash, Sigma-Delta, successive approximation register (SAR), etc. Apipelined, or sub-ranging, ADC uses two or more steps of sub-ranging. Acoarse conversion of an analog input voltage to a coarse digital valueis done, then the coarse digital value is converted back to an analogsignal with a digital-to-analog converter (DAC). The coarse analog valueis compared to the input voltage with an analog comparator, and thedifference, or residue, is then converted into a finer digitalrepresentation and the results are combined.

BRIEF DESCRIPTION OF THE DRAWINGS

Aspects of the present disclosure are best understood from the followingdetailed description when read with the accompanying figures. It isnoted that, in accordance with the standard practice in the industry,various features are not drawn to scale. In fact, the dimensions of thevarious features may be arbitrarily increased or reduced for clarity ofdiscussion.

FIG. 1A is a block diagram illustrating aspects of an analog-to-digitalconverter (ADC) system in accordance with some embodiments.

FIG. 1B is a circuit diagram illustrating further aspects of ananalog-to-digital converter (ADC) in accordance with some embodiments.

FIG. 2 is a phase diagram illustrating the ADC clock signals inaccordance with some embodiments.

FIG. 3 is a circuit diagram illustrating aspects of a current steeringdigital-to-analog conversion (DAC) and transconductance (Gm) stage of anADC in accordance with some embodiments.

FIGS. 4 A and B are circuit diagrams depicting aspects of a residueamplifier of an ADC in accordance with some embodiments.

FIG. 5 is a circuit diagram illustrating aspects of a successiveapproximation register (SAR) stage of an ADC in accordance with someembodiments.

FIG. 6 is a phase diagram illustrating the SAR clock signals inaccordance with some embodiments.

FIG. 7 depicts an analog-to-digital conversion method in accordance withsome embodiments.

DETAILED DESCRIPTION

The following disclosure provides many different embodiments, orexamples, for implementing different features of the provided subjectmatter. Specific examples of components and arrangements are describedbelow to simplify the present disclosure. These are, of course, merelyexamples and are not intended to be limiting. For example, the formationof a first feature over or on a second feature in the description thatfollows may include embodiments in which the first and second featuresare formed in direct contact, and may also include embodiments in whichadditional features may be formed between the first and second features,such that the first and second features may not be in direct contact. Inaddition, the present disclosure may repeat reference numerals and/orletters in the various examples. This repetition is for the purpose ofsimplicity and clarity and does not in itself dictate a relationshipbetween the various embodiments and/or configurations discussed.

Further, spatially relative terms, such as “beneath,” “below,” “lower,”“above,” “upper” and the like, may be used herein for ease ofdescription to describe one element or feature's relationship to anotherelement(s) or feature(s) as illustrated in the figures. The spatiallyrelative terms are intended to encompass different orientations of thedevice in use or operation in addition to the orientation depicted inthe figures. The apparatus may be otherwise oriented (rotated 90 degreesor at other orientations) and the spatially relative descriptors usedherein may likewise be interpreted accordingly.

Analog-to-digital converters (ADC) convert an analog voltage signal intoa digital signal. For example, a pipelined, or sub-ranging, ADC uses twoor more steps of sub-ranging. A coarse conversion of an analog inputvoltage to a coarse digital value is done, then the coarse digital valueis converted back to an analog signal with a digital-to-analog converter(DAC). The coarse value is compared to the input voltage with an analogcomparator, and the difference, or residue, is then converted finer andthe results are combined.

A successive-approximation ADC uses a comparator to successively narrowa range that contains the input voltage. At each successive step, theconverter compares the input voltage to the output of a DAC that mightrepresent the midpoint of a selected voltage range. At each step in thisprocess, the approximation is stored in a successive approximationregister (SAR). The steps are continued until the desired resolution isreached. With some ADC methods, it can be difficult to attain asufficiently high signal to noise ratio (SNR) and conversion bandwidthin low voltage deep submicron processes.

Some pipelined ADC methods use a switched capacitor Multiplying DAC(MDAC) which tends to be limited in conversion bandwidth. Whilepipelined ADCs can provide high resolution and high bandwidthconversion, they also tend to be power hungry because they use severalswitched capacitor MDACs. Similarly, while SAR ADCs provide a relativelylow power architecture, they also use a traditional switched capacitorMDAC. Such ADC methods may not be readily scalable to deep sub-micronprocess technologies while attaining good power efficiency.

The present disclosure includes examples of a multi-stage pipelined ADCwith a current steering first stage and a cascaded SAR second stage. Asdiscussed in further detail below, disclosed examples employ bothcurrent domain and voltage domain signal processing to attain highsampling rates and low power consumption. In some disclosed examples,this is achieved by the use of a low power current steering DACapproach, where a combined current steering DAC and a transconductanceamplifier cell are employed. A high conversion rate is achievablebecause the proposed current steering DAC is inherently faster than aswitched capacitor method for the same power consumption by essentiallyreplacing the switched capacitor network of a conventional switchedcapacitor MDAC with feedback resistors to convert the residue currentsignals to a voltage signal.

FIG. 1A generally illustrates an example of an ADC system 100 inaccordance with some disclosed embodiments. In general, the ADC system100 includes a first MDAC stage 10 coupled to an input terminal 102 thatreceives an analog input voltage signal V_(IP)/V_(IM). The first MDACstage 10 includes a first sub-ADC stage 30 configured to output a firstdigital value corresponding to the analog input voltage. In someexamples, the first digital value output by the first MDAC stage 10 isthe most significant bits (MSB) of the ADC digital output signal. Thefirst MDAC stage 10 further includes a current steering DAC stage 40that is connected to the input terminal 102 and receives the output ofthe first MDAC stage 30. The current steering DAC stage 40 converts theanalog input voltage and the first digital value to respective first andsecond current signals, determines a residue current signal representinga difference between the first current signal and the second currentsignal in the current domain, and converts the residue current signal toan analog residual voltage output signal Vres.

A second ADC stage 20 is coupled to the first MDAC stage 10 to receivethe analog residual voltage signal Vres, and convert the analog residuevoltage signal Vres to a second digital value, which in the illustratedexample is the least significant bits (LSB) of the ADC digital outputsignal. An alignment and digital error correction stage 50 is configuredto combine the first and the second digital values MSB, LSB and output adigital value Dons representing the analog input signals at an outputterminal 104.

FIG. 1B depicts further aspects of the ADC system 100 shown in FIG. 1A.In FIG. 1B, the first sub ADC stage 30 of the first MDAC stage 10includes a flash sub-ADC 125, while the second ADC stage 20 includes acascaded successive approximation register (SAR) 135 ADC.

The input terminal 102 is configured to receive differential analoginput signals V_(IP) and V_(IM), which are sampled by a switchedcapacitor network 185. As discussed further below, various controlsignals (170, 175, 180) are provided to control the operation of aplurality of switches 101 as well as the flash ADC 125 and the currentsteering digital-to-analog converter (IDAC) 130.

The illustrated first sub-ADC stage 30 includes a flash sub-ADC 125 togenerate the first digital value MSB of the digital output signalV_(out). The flash sub-ADC 125 receives the differential analog inputsignals V_(IP), V_(IM) and converts this analog signal to the firstdigital value at an output terminal that is connected to the alignmentand digital error correction stage 50 and the current steering DAC stage40. The current steering DAC stage 40 is comprised of a transconductanceamplifier (Gm) 115 to perform a voltage-to-current conversion of thesampled input signal. In the illustrated example, the Gm 115 does notreceive a current feedback signal and thus operates open loop. Examplesof the current steering DAC stage 40 also have a current steering IDAC130 configured to receive and convert the first digital value receivedfrom the flash sub-ADC 125 back to an analog representation in thecurrent domain. The Gm 115 and IDAC 130 output currents are combined togenerate a residue current representing the difference between the firstdigital value and the input voltage, which is then converted to thevoltage residue signal Vres and output to the second ADC stage 20. Asdiscussed further below, in some examples, the operations of the Gm cell115 and the IDAC 130 are merged or combined into a common circuit, thussimplifying the actual circuit implementation. Additionally, theGain-Bandwidth requirements of the residue amplifier may besignificantly reduced since the Gain-Bandwidth is inversely proportionalto the residue amplifier feedback factor and the disclosed amplifier hasa feedback factor close to unity. This in turn reduces the powerconsumption of the circuit compared to conventional switched capacitorMDAC methods.

The residue amplifier 120 is configured to receive the residue currentIres,p/Ires,m from the Gm 115 and IDAC 130, and convert the residuecurrent Ires,p/Ires,m into the residue voltage signal Vres based on thefeedback resistors 190. The residue voltage Vres represents thedifference between the analog input voltage and the first digitalrepresentation of the analog input voltage signal output by the Flashsub-ADC 125. As discussed further herein below, the residue amplifiermay include two stages, the first stage employing a wideband self-biasedamplifier and the second stage having a common mode feedback circuit.The residue voltage is then passed to the second ADC stage 20 of the ADC100. Employing a current mode processing of the residue, rather than aswitched-capacitor device reduces influences of capacitor loading on theresidue amplifier to improve performance.

In the illustrated example, the second ADC stage 20 is coupled toreceive the residue voltage output by the current steering DAC stage 40.Additionally, second ADC stage 20 is configured to convert the residuevoltage into the second digital value representing the least significantbits (LSBs) of the digital output signal. The MSBs and LSBs are receivedand combined in the alignment and digital error correction stage 50,which outputs the digital representation D_(out) of the analog inputvoltage at the output terminal 104. In the illustrated example, thefirst ADC stage 10 provides 4 bit MSBs, and the second ADC stage 20provides 9 bit LSBs to the alignment and digital error correction stage,which provides a 12 bit digital output signal (one bit is redundant andis used to accomplish the digital error correction function).

Using current steering instead of switch capacitors in the first MDACstage 10 reduces the gain-bandwidth (GBW) requirements of the amplifierfor a given settling accuracy. Referring to Equations 1 and 2 shownbelow, generally, the gain bandwidth is required to be greater than orequal to two times the sampling frequency (F_(S)) times the natural logof two times the number of bits converted after the first MDAC stage 10,or “backend bits” (N_(BACKEND)) all over the feedback factor (β) (seeEquation 1). In Equations 1 and 2, β is defined as R_(GM) (or R_(DAC))divided by R_(GM) (or R_(DAC)) plus R_(F), see Equation 2. The feedbackfactor β is close to unity in the current steering DAC since theresistance of R_(F) 190 is small compared to the resistance of the IDAC130 and GM 115. A switched capacitor MDAC has a β much less than unity,which indicates a larger GBW is required for the same settling accuracyvis-à-vis the current steering approach.

$\begin{matrix}{{GBW} \geq \frac{2*F_{s}*{{Ln}(2)}*N_{backend}}{\beta}} & (1) \\{\beta = {\frac{\left. R_{GM}||R_{DAC} \right.}{\left. R_{GM}||{R_{DAC} + R_{F}} \right.} \approx 1}} & (2)\end{matrix}$

Thus, the Gain-Bandwidth requirements of the disclosed residue amplifierare significantly reduced, since the Gain-Bandwidth is inverselyproportional to the feedback factor, β, and the example residueamplifier arrangement has a feedback factor close to unity. As anexample, GBW requirements for a 12 bit/500MSPS ADC using currentsteering would be less than 10 GHz, while an ADC of the samespecifications using switched capacitors instead of current steeringwould have a GBW requirement of greater than 100 GHz.

FIG. 2 depicts an example of the control signals 170, 175 for theswitches 101, the Gm 115, the flash ADC 125, the IDAC 130, the residueamplifier 120, and the SAR 135. The switches 101 turn on when the firstphase control signal 170 goes high at a time 230, resulting in thesampling capacitors 185 charging based on the analog input signalsV_(IP), V_(IM). When the first phase signal 170 goes low at time 232,the switches 170 open and the Gm 115 and the flash ADC 125 sample theanalog input signals V_(IP), V_(IM). The flash ADC 125 and the IDAC 130complete the respective conversions and latches the output based on thesecond phase signal 175 going high at a time 220. Additionally, when thefirst phase signal 170 is high, the residue current signals from the Gm115 and IDAC 130 are not output to the residue amplifier 120 while theinput signals V_(IP), V_(IM) are sampled, allowing the residue amplifier120 to be reset.

FIG. 3 is a circuit diagram 300 illustrating aspects of an example ofthe IDAC 130 and transconductance amplifier (Gm) 115 of the ADC 100. Insome examples, the IDAC 130 is comprised of a plurality of DAC unitcells 130, and the Gm 115 may also include a plurality of Gm cells 115.

Each of the IDAC 130 cells includes transistors 340 and 342. Thetransistor 340 is controlled by bias voltage V_(b1) 315 and is connectedbetween a voltage terminal V_(DD) and the transistor 342, which receivesa bias voltage V_(b2) at its gate terminal. The transistors 340 and 342are configured to provide a current source representing the leastsignificant digits I_(LSb1) to transistors 344 and 346 based on the MSBdigital output signal. Control signals D and DZ are received atrespective gate terminals to control the operation of the transistors344 and 346 to output the residual current signals I_(resp), I_(resm)representing the analog residual voltage signal. The control signals Dand DZ are provided by the first stage Flash sub-ADC 125 outputs.

The Gm cell 115 converts the analog input voltage signals V_(IP) andV_(IM) from the voltage domain into a representation of the voltage inthe current domain. The GM cell 115 includes current sources 355 andresistors R_(S) 360. The transistors 348, 350 are connected to thecurrent sources 355, with the analog input voltage signals V_(IP) andV_(IM) coupled to the respective gate terminals of the transistors 348,350. The Gm cell 115 thus provides current signals Igm,p/Igm,mrepresenting the sampled analog input voltage signals V_(IP) and V_(IM).As noted above, the IDAC unit cell 130 outputs a current signalIdac,p/Idac,m representing the analog input signals. The IDAC 130 and Gm115 output the residue current signals Ires,p/Iresm, which representsthe difference between the current signals Igm,p/Igm,m output by the Gmunit cell 115 and the IDAC cells 130 Idac,p/Idac,m. The residue currentsignal Idac,p/Idac,m is received by the residue amplifier 120, whichoutputs the voltage residue signals that represents the differencebetween the sample input voltage and the first digital signal output bythe sub ADC 125.

Thus, the operations of the Gm 115 and the IDAC 130 may be merged into acommon circuit as shown in FIG. 3, which may simplify the circuitimplementation. More particularly, in some disclosed examples theincoming voltage signals are converted into current, avoltage-to-current converter (Gm cell 115) may be used to perform thetransformation between the voltage and current domains. The Gm cell 115yields a current given by the Equation 3 shown below.

$\begin{matrix}{{I_{resp} - I_{resm}} = {{1I_{LSB}} - \frac{\left( {V_{IP} - V_{IM}} \right)}{2R_{S}} - \left( {{2I_{B}} + I_{C}} \right)}} & (3)\end{matrix}$

As seen from Equation 3, the output current (I_(resp)−I_(resm)) of thecombined circuit yields a current containing the IDAC 130 current(1I_(LSB)), the Gm 115 current (V_(IP)−V_(IM) divided by 2R_(S)) andbias currents (2I_(B)+I_(C)), and therefore represents a “merged”current comprising the Gm and DAC currents.

FIGS. 4A and 4B illustrate aspects of an example of the residueamplifier 120, which may be a transimpedance amplifier configured toconvert the residue current signal to an analog voltage signal output tothe second ADC stage 20. In some examples, the residue amplifier 120 isa two-stage residue amplifier with feed forward compensation.

The residue amplifier 120 illustrated in FIG. 4A may be a fullydifferential amplifier that comprises a first stage 402 and a secondstage 404. The first stage 402 contains input pair transistors 440 b and440 c having gate terminals coupled to the differential input voltagesignals V_(IP)/V_(IM), with a load that may include a resistor pair 420,and transistors 415 b and 415 c. The bias current is set by a currentsource 445 including the residue current Ires output by the combined Gm115 and IDAC 130. The differential outputs Vres,p/Vres,m of the firststage 402 represent the residue signal output by the first MDAC stage10, and are coupled to the inputs of the second stage 404 viatransistors 415 d and 415 a. The second stage 404 includes transistors440 d, 415 d, 440 a and 415 a. The resistors 190 feedback the residuevoltage signals Vres,p and Vres,m from the second-stage output back tothe first stage inputs V_(IN) and V_(IP). The amplifier is compensatedby capacitors 435 and 430.

The resultant currents provided by the Gm 115 and IDAC 130 are amplifiedand converted back into the voltage domain for later use by the SAR 135to produce the LSBs of the digital output signal Dout. The differentialinput voltages V_(IM) and V_(IP), are coupled to the current signals Igmand Idac via the transistors 350 and 348 as shown in FIG. 3. Inaddition, the residue signals Vres,p/Vres,m output by the amplifier areconnected to the input voltages V_(IN) and V_(IP) in a negative feedbackconfiguration through the feedback resistors 190.

FIG. 4B shows an example of a common-mode feedback circuit 410 thatprovides a common mode feedback voltage signal V_(CMFB) to the amplifier400 shown in FIG. 4A. The residue voltage signals Vres,p/Vres,m outputby the amplifier 400 are sampled by a common mode detector circuit 472.The common mode detector 472 is comprised of capacitors 480 andresistors 475. The common mode detector 472 takes an average of thesignals Vres,p, Vres,m and that average is sent to the positive terminalof an error amplifier 490. The error amplifier 490 compares the averagedvoltage to the common mode voltage V_(CM) and outputs the common modefeedback voltage V_(CMFB) to the first stage of the residue amplifier400 via the resistors 465.

FIG. 5 depicts an example of the second ADC stage 20, which includes aSAR ADC 135 in some examples. The SAR ADC 135 includes a sample and holdcircuit 501, a comparator 560 and SAR logic 140. The SAR logic 140receives a clock signal CLKS and provides an output signal ϕ_(SAR) tothe sample and hold circuit 501. The output signal ϕ_(SAR) controls theoperation of a plurality of switches 510 to selectively connect one sideof a plurality of capacitors. In the illustrated example, there are twosets of capacitors 520 a, 520 b corresponding respectively to thedifferential residue voltage inputs Vres,p and Vres,m. Each of the setsof capacitors 520 a, 520 b includes a plurality of capacitors C₀-C_(N),where N may correspond to the number of bits to be converted, such asthe LSB bits shown in FIG. 1B. In some examples, the capacitors arebinary weighted, with the minimum capacitor size C being about 2fF insome embodiments. The control signal 170 controls various SPDT switches505 used to connect the sample and hold circuit 501 to the input residuevoltages Vres,p and Vres,m, and to a reference signal V_(ref) as well asswitches 101 connecting the common mode voltage signal V_(CM) to thesample and hold circuit 501 and the comparator 560.

Referring to FIG. 2, when the control signal ϕ₁ 170 is high, switches505 connect the analog residue input signals Vres,p and Vres,m to thetop plates of the capacitors C₀-C_(N) of the sets of capacitors 520 a,520 b. At the same time, the bottom plates of the capacitors C₀-C_(N)and the inputs of the comparator 560 are coupled to the common modevoltage V_(CM) due to switches 101 closing. During the next phase theoutput signal ϕ_(SAR) pulses are asserted to control the binary searchalgorithm implemented by the SAR logic 135 to generate the seconddigital output representing the LSBs of the analog input voltage.

FIG. 6 illustrates an example phase diagram for the control signal ϕ₁170 and the output signal ϕ_(SAR). The control ϕ₁ 170 is used fortracking and holding the signal in the SAR ADC 135 of the second ADCstage 20 of the ADC 100 while the output signal ϕ_(SAR) is used tocontrol the operation of the switches 510 shown in FIG. 5. When theinput signal ϕ₁ goes low as shown at a time 602, the output signalϕ_(SAR) starts cycling and provides a plurality of output pulses 604 tocontrol the switches 510 and sample the differential input signalsV_(IP) and V_(IM) via the sets of capacitors 520 a, 520 b. When thecontrol signal ϕ₁ 170 goes high such as at the time 606, the outputsignal ϕ_(SAR) stops cycling and the output pulses 604 cease.

FIG. 7 shows an example of an ADC method 700 implemented by the ADC 100.The method 700 starts at block 702 where an analog input voltage V_(IP),V_(IM) is received, for example, at the input terminals 102 shown inFIG. 1B. At block 704 the analog input voltage is converted to a firstdigital value, which may be the MSBs for the digital output signal, andat block 706 the analog input voltage is converted into a first currentI_(gm,p), I_(gm,m). At block 708 the first digital value is converted toa second current I_(dac,p), I_(dac,m). The first and second currentsignals are combed into a residue current signal I_(res,p), I_(res,m) inblock 710. At block 712, and analog residue voltage is generated fromthe residue current. This residue voltage is then used in the generationof a second digital value in block 714, which may be the LSBs of thedigital output signal. In block 716 the first digital value and thesecond digital value are combined to create the digital output signalrepresenting the analog input signal.

Accordingly, the various embodiments disclosed herein provide an ADCmethod and system that can achieve a high conversion rate and highaccuracy with good power efficiency. Disclosed embodiments include afirst ADC stage with a first sub-ADC stage configured to output a firstdigital value corresponding to an analog input voltage. A currentsteering DAC stage is configured to convert the analog input voltage andthe first digital value to respective first and second current signals,determine a residue current signal representing a difference between thefirst current signal and the second current signal in the currentdomain, and convert the residue current signal to an analog residualvoltage signal. A second ADC stage is coupled to the first ADC stage toreceive the analog residual voltage signal, and convert the analogresidue voltage signal to a second digital value. An alignment anddigital error correction stage is configured to combine the first andthe second digital values into a digital output voltage.

In accordance with additional embodiments, an ADC conversion methodincludes receiving an analog input voltage signal, converting the analoginput voltage signal to a first digital signal, and converting theanalog input voltage to a first current signal. The first digital valueis converted to a second current signal, and the first and secondcurrents are combined into a residue current signal. The residue currentsignal is converted to an analog residue voltage signal, and the analogresidue voltage signal is converted to a second digital signal. Thefirst and second digital signals are combined into a digital outputsignal representing the analog input voltage signal.

In accordance with still further examples, an ADC has an input terminalconfigured to receive an analog input voltage. A sub ADC is configuredto sample the received analog input voltage signal and output a firstdigital signal representing the analog input voltage signal. Atransconductance amplifier is configured to sample the received analoginput voltage signal and output a first current signal. A DAC converteris configured to receive the first digital signal and output a secondcurrent signal representing the first digital signal, and a residualamplifier is configured to receive the first and second current signalsand output an analog residual voltage signal based on the first andsecond current signals. A residue ADC is configured to receive theanalog residue voltage signal and output a second digital signalrepresenting the analog residue voltage signal. An alignment and errorcorrection circuit is configured to combine the first and second digitalsignals.

This disclosure outlines various embodiments so that those skilled inthe art may better understand the aspects of the present disclosure.Those skilled in the art should appreciate that they may readily use thepresent disclosure as a basis for designing or modifying other processesand structures for carrying out the same purposes and/or achieving thesame advantages of the embodiments introduced herein. Those skilled inthe art should also realize that such equivalent constructions do notdepart from the spirit and scope of the present disclosure, and thatthey may make various changes, substitutions, and alterations hereinwithout departing from the spirit and scope of the present disclosure.

What is claimed is:
 1. An analog-to-digital converter (ADC), comprising:an input terminal configured to receive an analog input voltage; a firstMDAC stage including a first sub-ADC stage coupled to the input terminaland configured to output a first digital value corresponding to theanalog input voltage, and a current steering DAC stage connected to theinput terminal and the output terminal of the first sub-ADC stage, thecurrent steering DAC stage including a transconductance amplifier (Gm)and a current steering digital to analog converter (DAC), thetransconductance amplifier configured to convert the analog inputvoltage to a first current signal, the current steering DAC configuredto convert the first digital value to a second current signal, thecurrent steering DAC stage configured to determine a residue currentsignal representing a difference between the first current signal andthe second current signal in the current domain, and convert the residuecurrent signal to an analog residual voltage signal, wherein thetransconductance amplifier includes a first plurality oftransconductance amplifier cells, and the current steering DAC includesa second plurality of current steering DAC unit cells; a second ADCstage coupled to the current steering DAC stage to receive the analogresidual voltage signal, and convert the analog residue voltage signalto a second digital value; an alignment and digital error correctionstage configured to combine the first and the second digital values intoa digital output voltage; and an output terminal coupled to thealignment and error correction stage configured to output the digitaloutput voltage.
 2. The ADC of claim 1, wherein the first sub-ADC stageincludes a flash ADC coupled to the input terminal configured to convertthe analog input voltage to the first digital value.
 3. The ADC of claim1, wherein the current steering DAC stage is configured to combine thefirst current signal and the second current signal to generate theresidue current signal in the current domain.
 4. The ADC of claim 1,wherein the current steering DAC stage includes a residue amplifierconfigured to convert the residue current signal to the analog residuevoltage signal.
 5. The ADC of claim 4, wherein the residue amplifierincludes first and second stages, the first stage having a widebandself-biased amplifier and the second stage including a common modefeedback circuit.
 6. The ADC of claim 1, wherein the second ADC stageincludes a successive approximation register (“SAR”) ADC.
 7. The ADC ofclaim 1, wherein the transconductance amplifier and the current steeringDAC form an output node at a junction of the transconductance amplifierand the current steering DAC; wherein each current steering DAC unitcell includes a voltage terminal, a first transistor, a secondtransistor, a first pair of differential transistors including a thirdtransistor and a fourth transistor, the first transistor connectedbetween the voltage terminal and the second transistor, the secondtransistor connected between the first transistor and the first pair ofdifferential transistors, gates of the first transistor and the secondtransistor receive bias voltages to provide a current source, and gatesof the third transistor and the fourth transistor receive controlsignals to output the second current signal at the output node; andwherein each transconductance amplifier cell includes a third pluralityof current sources, a fourth plurality of resistors, a second pair ofdifferential transistors including a fifth transistor and a sixthtransistor, and gates of the fifth transistor and the sixth transistorreceive the analog input voltage to output the first current signal atthe output node.
 8. An analog-to-digital (ADC) conversion method,comprising: receiving an analog input voltage signal; converting theanalog input voltage signal to a first digital signal by a sub ADC,wherein the sub ADC is configured to sample the received analog inputvoltage signal in response to a first phase clock signal, and the outputthe first digital signal in response to a second phase clock signal;converting the analog input voltage to a first current signal by atransconductance amplifier (Gm), wherein the Gm is configured to samplethe received analog input voltage signal in response to the first phaseclock signal; converting the first digital value to a second currentsignal by a current steering digital to analog converter (DAC);determining a difference between the first and second current signals todetermine a residue current signal; converting the residue currentsignal to an analog residue voltage signal; converting the analogresidue voltage signal to a second digital signal; and combining thefirst and second digital signals into a digital output signalrepresenting the analog input voltage signal.
 9. The analog-to-digitalconversion method of claim 8, further comprising generating the residuevoltage using a common mode feedback loop.
 10. The analog-to-digitalconversion method of claim 8, further comprising converting the analoginput voltage to the first digital value using a flash ADC.
 11. Theanalog-to-digital conversion method of claim 10, further comprisingdetermining the difference between the first and second current signalsat a summing junction.
 12. An analog-to-digital converter (ADC),comprising: an input terminal configured to receive an analog inputvoltage; a sub ADC configured to sample the received analog inputvoltage signal and output a first digital signal representing the analoginput voltage signal; a transconductance amplifier configured to samplethe received analog input voltage signal and output a first currentsignal wherein the transconductance amplifier includes a first pluralityof transconductance amplifier cells; a current steering digital toanalog converter (DAC) configured to receive the first digital signaland output a second current signal representing the first digitalsignal, wherein the current steering DAC includes a second plurality ofcurrent steering DAC unit cells; a residual amplifier configured toreceive the first and second current signals and output an analogresidual voltage signal based on a difference between_the first andsecond current signals; a residue ADC configured to receive the analogresidue voltage signal and output a second digital signal representingthe analog residue voltage signal; and an alignment and error correctioncircuit configured to combine the first and second digital signals. 13.The ADC of claim 12, wherein the sub ADC includes a flash ADC.
 14. TheADC of claim 12, wherein the sub ADC is configured to sample thereceived analog input voltage signal in response to a first phase clocksignal.
 15. The ADC of claim 14, wherein the transconductance amplifieris configured to sample the received analog input voltage signal inresponse to the first phase clock signal.
 16. The ADC of claim 15,wherein the sub ADC is configured to output the first digital signal inresponse to a second phase clock signal.
 17. The ADC of claim 16,wherein the current steering DAC is configured to output the secondcurrent signal in response to the second phase clock signal.
 18. The ADCof claim 15, wherein the residual amplifier is configured to reset inresponse to the first phase clock signal.
 19. The ADC of claim 12,wherein the residual amplifier is configured to amplify the differencebetween the first and second current signals.
 20. The ADC of claim 12,wherein the transconductance amplifier and the current steering DAC forman output node at a junction of the transconductance amplifier and thecurrent steering DAC; wherein each current steering DAC unit cellincludes a voltage terminal, a first transistor, a second transistor, afirst pair of differential transistors including a third transistor anda fourth transistor, the first transistor connected between the voltageterminal and the second transistor, the second transistor connectedbetween the first transistor and the first pair of differentialtransistors, gates of the first transistor and the second transistorreceive bias voltages to provide a current source, and gates of thethird transistor and the fourth transistor receive control signals tooutput the second current signal at the output node; and wherein eachtransconductance amplifier cell includes a third plurality of currentsources, a fourth plurality of resistors, a second pair of differentialtransistors including a fifth transistor and a sixth transistor, andgates of the fifth transistor and the sixth transistor receive theanalog input voltage to output the first current signal at the outputnode.